Receiver second order intermodulation correction system and method

ABSTRACT

A system for correcting a second order intermodulation product in a direct conversion receiver is provided. The system includes a cross-covariance system receiving a data signal and a second order intermodulation estimate signal and generating a cross-covariance value. An auto-covariance system receives the second order intermodulation estimate signal and generates an auto covariance value. A buffer system stores a second order intermodulation product correction factor. A divider receives the cross-covariance value, the auto-covariance value and the second order intermodulation product correction factor and generates a running average second order intermodulation product correction factor.

The present application is a continuation of U.S. Pat. No. 8,032,102,issued Oct. 4, 2011, entitled “RECEIVER SECOND ORDER INTERMODULATIONCORRECTION SYSTEM AND METHOD” which is hereby incorporated by referencefor all purposes.

FIELD OF THE INVENTION

The present invention relates to radio frequency receivers, and moreparticularly to a receiver second order intermodulation productcorrection system and method, such as for use with a direct conversionreceiver.

BACKGROUND OF THE INVENTION

In a direct-conversion receiver, second-order distortion can occur thatresults in demodulation of the amplitude of the transmitted signal andgenerates unwanted signals in the baseband. This distortion becomes aproblem if a strong out-of-band blocker is present when the strength ofthe desired signals is small, such that the receiver is set with largegain. While SAW filters can be used to remove such distortion, SAWfilters are large and are generally implemented as off-chip modules,which increases the size and complexity of a receiver.

SUMMARY OF THE INVENTION

Therefore, a receiver second order intermodulation product correctionsystem and method are provided that eliminate the need for off-chipfilters. In particular, an estimate of the second order intermodulationproduct for a current transmission slot is used to update a second orderintermodulation product correction factor for a subsequent transmissionslot.

In accordance with an exemplary embodiment of the invention, a systemfor correcting a second order intermodulation product in a directconversion receiver is provided. The system includes a cross-covariancesystem receiving a data signal and a second order intermodulationestimate signal and generating a cross-covariance value. Anauto-covariance system receives the second order intermodulationestimate signal and generates an auto covariance value. A buffer systemstores a second order intermodulation product correction factor. Adivider receives the cross-covariance value, the auto-covariance valueand the second order intermodulation product correction factor andgenerates a running average second order intermodulation productcorrection factor.

The present invention provides many important technical advantages. Oneimportant technical advantage of the present invention is a second orderintermodulation product correction system and method that eliminates theneed for off-chip filters.

Those skilled in the art will further appreciate the advantages andsuperior features of the invention together with other important aspectsthereof on reading the detailed description that follows in conjunctionwith the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of a system for a receiver with second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention;

FIG. 2 is a diagram of a system for performing second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention;

FIG. 3 is a diagram of a system for generating a compensated direct andquadrature phase output to compensate for a second order intermodulationproduct in accordance with an exemplary embodiment present invention;and

FIG. 4 is a flowchart of a method for applying second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

In the description which follows, like parts are marked throughout thespecification and drawing with the same reference numerals,respectively. The drawing figures may not be to scale and certaincomponents may be shown in generalized or schematic form and identifiedby commercial designations in the interest of clarity and conciseness.

FIG. 1 is a diagram of a system 100 for a receiver with second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention. In one exemplary embodiment, system100 can be used in direct conversion systems, so as to remove secondorder intermodulation product effects created by blocker signals byusing a running estimate of the correction factor for successivetransmission slots, or in other suitable applications.

System 100 includes antennae 102, which receives a transmitted signal.The transmitted signal is amplified by low noise amplifier (LNA) 104 andis provided to mixer 106 which extracts the baseband signal from thetransmitted signal, such as by mixing the received signal with a localoscillator signal LO.

The baseband signal output by mixer 106 is provided to filter 108, whichcan be an analog infinite impulse response (IIR) filter or othersuitable filters, and adaptive gain unit 110, which provide DC offsetcorrection. In one exemplary embodiment, in a direct conversionreceiver, the large gain of the analog baseband can cause small DCoffsets that can saturate subsequent stages and substantially increasethe dynamic range requirements for analog to digital conversion.Periodic DC offset correction can be performed in the analog domain byaccumulating the output of a sigma-delta analog to digital converterbefore the received slot begins, to get an estimate of the DC offset.This offset can then be subtracted in a current mode at the output offilter 108. This procedure can be repeated after a predeterminedsettling time to the output of filter 112.

Filter 112, which can be an analog IIR filter or other suitable filters,and adaptive gain unit 114 form a bandwidth calibration stage. Processand temperature variations can cause frequencies to vary up toapproximately plus or minus 20% due to variations in sheet resistance,capacity, density or other variables. Therefore, periodic calibrationcan be performed to ensure that the bandwidth of filter 112 is correct.In one exemplary embodiment, calibration can be implemented by switchingcapacitors in or out in a binary weighted capacitor bank, or in othersuitable manners.

The offset corrected and bandwidth calibrated signal is provided toanalog to digital converter 116, which converts the analog signal to adigital signal having a suitable word size. The output from analog todigital converter is then provided to correlator 124 and summation unit128. In one exemplary embodiment, other suitable signal processing, suchas direct and quadrature phase signal processing, can also oralternatively be utilized.

The output from low noise amplifier 104 is also provided to envelopedetector 118, which generates a peak envelope signal. This signal isprovided to analog to digital converter 120, and is then output tomultiplier 122 which squares the output of the detected envelope signal.This output is provided to correlator 124 and adaptive gain unit 126.

Correlator 124 receives the intermodulation estimate from multiplier 122and correlates the estimate to the received signal. The output ofcorrelator 124 is provided to adaptive gain unit 126 where an adaptivegain factor is applied in order to remove the intermodulation component.

In one exemplary embodiment, the effect of the second-orderintermodulation product (IP2) on the received signal can be corrected bycalculating a running average of a second-order distortion coefficienta₂ for both the direct and quadrature phase components, which isestimated and updated at the end of every received transmission slot.The newly updated a₂ can then be used to compensate the next slot'ssecond-order distortion. The following exemplary moving average methodcan be used to calculate the new value for a₂:a ₂(k+1)=λ·a ₂(k)+(1−λ)·a ₂ _(—) _(est)(k)where a₂ _(—) _(est) is the coefficient estimation of the current slot,k is the slot index and λ is an empirically-determined value between 0and 1, but typically having a value of 0.5. Likewise, other suitablemethods can be used to calculate values for a₂.

IP2 correction can be performed using the auto-covariance of theenvelope signal s(t) and the cross-covariance between the receiveddirect or quadrature phase signal {tilde over (r)}(t) and s(t) tocalculate a₂ _(—) _(est). In one exemplary embodiment, {tilde over(r)}(t) can be simplified as the summation of the desired signal {tildeover (x)}(t) and the second-order intermodulation product m(t)=a₂ _(—)_(est)·s(t). In this exemplary embodiment, both auto-covariance A andcross-covariance B produce only one sample in the case that both inputsare aligned:

$A = {{a\;{{cov}\left( {s(t)} \right)}} = {{\sum\limits_{t = 1}^{N}\left\lbrack {{s(t)} - \overset{\_}{s}} \right\rbrack^{2}} = {{{\sum\limits_{t = 1}^{N}{s^{2}(t)}} - {2\overset{\_}{s}{\sum\limits_{t = 1}^{N}{s(t)}}} + {N \cdot {\overset{\_}{s}}^{2}}} = {{\sum\limits_{t = 1}^{N}{s^{2}(t)}} - {N \cdot {\overset{\_}{s}}^{2}}}}}}$$B = {{x\;{{cov}\left( {{s(t)},{\overset{\sim}{r}(t)}} \right)}} = {{\sum\limits_{t = 1}^{N}{\left\lbrack {{s(t)} - \overset{\_}{s}} \right\rbrack \cdot \left\lbrack {{\overset{\sim}{r}(t)} - \overset{\overset{\_}{\sim}}{r}} \right\rbrack}} = {{{\sum\limits_{t = 1}^{N}{{s(t)}{\overset{\sim}{r}(t)}}} - {\overset{\overset{\_}{\sim}}{r}{\sum\limits_{t = 1}^{N}{s(t)}}} - {\overset{\_}{s}{\sum\limits_{t = 1}^{N}{\overset{\sim}{r}(t)}}} + {N \cdot \overset{\_}{s} \cdot \overset{\overset{\_}{\sim}}{r}}}\mspace{14mu} = {{\sum\limits_{t = 1}^{N}{{s(t)}{\overset{\sim}{r}(t)}}} - {N \cdot \overset{\_}{s} \cdot \overset{\overset{\_}{\sim}}{r}}}}}}$where N is the length of samples in one slot

Because {tilde over (r)}(t)={tilde over (x)}(t)+a₂ _(—) _(est)·s(t), theoutput of cross-covariance can be expressed as:

$\begin{matrix}{B = {{\sum\limits_{i = 1}^{N}{{s(t)}{\overset{\sim}{x}(t)}}} + {a_{2{\_ est}}{\sum\limits_{t = 1}^{N}{s^{2}(t)}}} - {N \cdot \left( {{\overset{\_}{s} \cdot \overset{\overset{\_}{\sim}}{x}} + {a_{2{\_ est}}{\overset{\_}{s}}^{2}}} \right)}}} \\{= {{a_{2{\_ est}}\left( {{\sum\limits_{t = 1}^{N}{s^{2}(t)}} - {N \cdot {\overset{\_}{s}}^{2}}} \right)} + {\sum\limits_{t = 1}^{N}{{s(t)}{\overset{\sim}{x}(t)}}} - {N \cdot \overset{\_}{s} \cdot \overset{\overset{\_}{\sim}}{x}}}}\end{matrix}$

Because the cross-covariance B between two unrelated signals is muchsmaller than the auto-covariance A of a signal, the gain estimate forthe signal is produced by dividing B with A:

$\frac{B}{A} = {{a_{2{\_ est}} + \frac{{\sum\limits_{t = 1}^{N}{s(t){\overset{\sim}{x}(t)}}} - {N \cdot \overset{\_}{s} \cdot \overset{\overset{\_}{\sim}}{x}}}{{\sum\limits_{t = 1}^{N}{s^{2}(t)}} - {N \cdot {\overset{\_}{s}}^{2}}}} = {{a_{2{\_ est}} + \frac{{x{cov}}\left( {{s(t)},{\overset{\sim}{x}(t)}} \right)}{a\;{{cov}\left( {s(t)} \right)}}} \cong a_{2{\_ est}}}}$

When the strength of a blocker signal is too small, IP2 correction canbe disabled by setting second-order distortion coefficient a₂ to zeroinstead of updating with estimated value. The decision of turning on/offcompensation can be done by comparing the auto-covariance of the blockersignal's amplitude with a predetermined threshold A_(thr). In oneexemplary embodiment, the following relationship can be used todetermine whether compensation should be disabled:

${a_{2}\left( {k + 1} \right)} = \left\{ \begin{matrix}{{{\lambda \cdot {a_{2}(k)}} + {\left( {1 - \lambda} \right) \cdot a_{2{\_ est}}}},} & {\frac{A}{R^{2}} \geq A_{thr}} \\{0,} & {\frac{A}{R^{2}} < A_{thr}}\end{matrix} \right.$

Lambda (λ) is a weighting factor that is used to adjust the weight givenin the running average to the current and previous gain estimates. Inone exemplary embodiment, lambda can have a value that is determinedbased on empirical analysis of the operating environment, or can beassigned a default value, such as 0.5. R is selected to match the gainin the receiver path. By dividing A with R² the comparison isindependent of the setting of the gain in the receiver path.

In operation, system 100 corrects a received signal to remove a secondorder intermodulation product. System 100 allows a direct conversionreceiver to be implemented without the use of SAW filters or otheroff-chip filters, and eliminates potential second order blocker signalsthrough the use of a second order intermodulation product correctionprocess. In this exemplary embodiment, a running average of the secondorder intermodulation product is derived from the output of an envelopedetector that processes the received signal, which is used to estimatethe amount of correction required to remove any potential blockersignals that may be present in the signal after processing by thereceiver.

FIG. 2 is a diagram of a system 200 for performing second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention. System 200 can be implemented inhardware, software or a suitable combination of hardware and softwareand can be one or more software systems operating on a digital signalprocessing platform or other suitable processing platforms. As usedherein, “hardware” can include a combination of discrete components, anintegrated circuit, an application-specific integrated circuit, a fieldprogrammable gate array, or other suitable hardware. As used herein,“software” can include one or more objects, agents, threads, lines ofcode, subroutines, separate software applications, two or more lines ofcode or other suitable software structures operating in two or moresoftware applications or on two or more processors, or other suitablesoftware structures. In one exemplary embodiment, software can includeone or more lines of code or other suitable software structuresoperating in a general purpose software application, such as anoperating system, and one or more lines of code or other suitablesoftware structures operating in a specific purpose softwareapplication.

System 200 includes second order cascaded integrator-comb (CIC)decimator 202. Second order CIC decimator 202 receives the receivedtransmitted signal after amplification by a suitable low noise amplifierand generates an envelope of the signal, which is used to estimate thepeak voltage of the received signal. The output of second order CICdecimator 202 is provided to multiplier 204, which squares the output.The squared output is then provided to filter stages 206 and 208, whichcan be IIR filters or other suitable filters, and which introduce adelay equal to the delay created by the receiver stage, so as to allowthe detected peak signal to be compared to the processed receivedsignal. The output of filters stages 206 and 208 is provided to downsampler 210, which down samples the data rate of the received signal soas to match the output from the receiver stage.

The output of down sampler 210 is then multiplied by a compensationfactor C by multiplier 212. In one exemplary embodiment, thecompensation factor is used to correct for mismatches between thereceiver path and the peak director path, such as from second order CICdecimator 202 gain difference and the extra gain in the receiver path.The output of multiplier 212 is then provided to cross covariancecalculators 214 and 216, auto covariance calculator 218, and mixers 228and 230.

Cross covariance calculators 214 and 216 generate a cross covarianceestimate for the received quadrature phase and in phase signals,respectively. Other suitable signal formats can also or alternatively beutilized. Auto covariance calculator 218 generates an auto covarianceestimate for the received envelope detector output. The output of thequadrature and direct phase covariance product from cross covariancecalculators 214 and 216, respectively, is provided to divider/buffers220 and 222, respectively, which divide the output of cross covariancecalculators 214 and 216 by the output from auto covariance calculator218, to generate a second order direct and quadrature phase output. Inaddition, divider/buffers 220 and 222 retrieve the gain values for theprevious slot and calculate a moving average by multiplying the previousslot gain value by a weighting factor lambda, and the current gain valueby (1−lambda), and adding the two gain values. The new gain value isbuffered for use in calculating the gain value for the next slot. Thegain-compensated direct and quadrature phase signals are then providedto multiplexers 224 and 226, respectively, which receive a control inputto operate as a switch, such as to generate a zero output if thestrength of the blocker signal is too small and second orderintermodulation correction is disabled. The output of multiplexers 224and 226 is multiplied with the detected envelope signal by mixers 228and 230, respectively. These outputs are then subtracted from thereceived direct and quadrature phase signals by subtractors 232 and 234,respectively, and the corrected direct and quadrature phase signals areoutput from subtractors 234 and 232, respectively.

In operation, system 200 performs second order intermodulation productcorrection on a direct and quadrature phase signal received from areceiver chain by using the envelope of a received signal to generateauto and cross covariance estimates. Likewise, the auto and crosscovariance estimates are stored for each successive slot and are updatedbased on the calculated estimate of the cross and auto covariance.

FIG. 3 is a diagram of a system 300 for generating a compensated directand quadrature phase output to compensate for a second orderintermodulation product, in accordance with an exemplary embodimentpresent invention. System 300 can be implemented in hardware, software,or a suitable combination of hardware and software and can be one ormore software systems operating on a digital signal processing platform.Likewise, other suitable received signals can also or alternatively beprocessed, such that system 300 is not limited to quadrature and directphase signal processing.

System 300 includes covariance calculator 302, which calculatescovariance based on output received from a peak detector of the receivedvoltage signal, as well as the direct and quadrature phase input from areceiver chain. Second order CIC decimator 304 receives a peak detectoroutput and generates a CIC peak detector output for input to covariancecalculator 302. Control logic 306 controls the operation of second orderCIC decimator 304 and dual order filter 308. Likewise, control logic 306provides a counter signal to covariance calculator 302. Covariancecalculator 302 outputs a squared peak value to delay unit 310, whichprovides a delay to the squared peak value so as to match the timing ofthe received direct and quadrature phase signals. The output of delayunit 310 is provided to dual order filter 308, which filters the squaredpeak signal so as to generate a time varying peak detector output tocovariance calculator 302. Covariance calculator 302 then uses the inputdirect and quadrature phase signals and the output of dual order filter308 to calculate a covariance and to compensate the input direct andquadrature phase signals. In one exemplary embodiment, the compensateddirect and quadrature phase signals can be equal to the input direct andquadrature phase signals, such as where it is determined that secondorder intermodulation product correction should not be applied, aspreviously discussed.

In operation, system 300 applies second order intermodulation productcorrection on received direct and quadrature phase signals, by applyinga running average of an estimated second order intermodulation productcorrection gain. Covariance calculator 302 can be implemented using adigital signal processor or other suitable platforms, so as to reducethe number of components required to implement the second orderintermodulation product correction factor calculation and application,such as calculation of cross-covariance, auto-covariance, compensationswitching based on the received signal strength, and other functions.

FIG. 4 is a flowchart of a method 400 for applying second orderintermodulation product correction in accordance with an exemplaryembodiment of the present invention. Method 400 can be used to calculatesecond order intermodulation correction factors for direct andquadrature phase signals in a direct conversion receiver, or for othersuitable processes.

Method 400 begins at 402 where the strength of a blocker signal isdetermined. In one exemplary embodiment, the strength of a blocker canbe determined based on the calculated value of the auto covariancedivided by a variable that is representative of the gain of the receiverpath. If the result of that division operation is greater than or equalto a predetermined setting, then second intermodulation productcorrection can be applied, whereas if it is below a predetermined limit,then second order intermodulation product correction can be disabled,such as where the strength of the block or signal is too small torequire second order intermodulation product correction. The method thenproceeds to 404.

At 404, it is determined whether to disable the second orderintermodulation product correction. If it is determined that secondorder intermodulation product correction should be applied, the methodproceeds to 406 where auto covariance of the peak detector envelopesignal is calculated, such as by using the previously described processor other suitable processes. The method then proceeds to 408 where crosscovariance is calculated between the peak detector envelope signal andthe received direct and quadrature phase signals, respectively, such asby using the previously described process or other suitable processes.Likewise, other suitable received signals can also or alternatively beused. The method then proceeds to 410 where a second order distortioncoefficient is calculated, such as by using the previously describedprocess or other suitable processes. The method then proceeds to 412where a running average for the second order distortion coefficient forthe direct and quadrature phase signals or other suitable signals iscalculated, such as by using the previously described process or othersuitable processes. The current value of the second order distortioncoefficient for the direct and quadrature phase signals or othersuitable signals is buffered for use in the next slot calculations, andthe method proceeds to 414 where the correction coefficient is appliedto the direct and quadrature phase signals or other suitable signals,such as by using the previously described process or other suitableprocesses. The method then proceeds to 416 where the slot isincremented. The method then returns to 402.

Likewise, if it is determined at 404 that second order distortioncorrection should be disabled, the method proceeds to 418 where thevalue for the second order distortion coefficient for the slot is set tozero and the running average is updated and buffered. The method thenproceeds to 420 where the slot is incremented, and the method returns to402.

In operation, the second order intermodulation product for a receivedsignal is corrected by calculating a running average of the second orderdistortion coefficient for the received signal, such as for direct andquadrature phase components or other suitable signals, which isestimated and updated at the end of every transmission slot using thesecond order intermodulation correction coefficient from the previousslot. In this manner, the effect of any blocker signals can be estimatedand used to control the second order intermodulation product, so as toeliminate the need for SAW filters in a direct conversion receiver.

In view of the above detailed description of the present invention andassociated drawings, other modifications and variations are apparent tothose skilled in the art. It is also apparent that such othermodifications and variations may be effected without departing from thespirit and scope of the present invention.

1. A system for correcting a second order intermodulation product in adirect conversion receiver, comprising a divider for receiving across-covariance value, an auto-covariance value and a second orderintermodulation product correction factor and for generating a runningaverage second order intermodulation product correction factor as afunction of the cross-covariance value, the auto-covariance value andthe second order intermodulation product correction factor.
 2. Thesystem of claim 1 further comprising a cross-covariance system forreceiving a data signal and a second order intermodulation estimatesignal and generating the cross-covariance value.
 3. The system of claim1 further comprising an auto-covariance system for receiving a secondorder intermodulation estimate signal and generating the auto covariancevalue.
 4. The system of claim 1 further comprising a buffer system forstoring the running average second order intermodulation productcorrection factor.
 5. The system of claim 1 further comprising amultiplier for multiplying a second order intermodulation estimatesignal and the running average second order intermodulation productcorrection factor to generate a correction signal.
 6. The system ofclaim 5 further comprising a subtractor for subtracting the correctionsignal from the data signal to generate a corrected data signal.
 7. Thesystem of claim 1 further comprising: a direct phase cross-covariancesystem receiving a direct phase data signal and a second orderintermodulation estimate signal and generating a direct phasecross-covariance value; and a quadrature phase cross-covariance systemreceiving a quadrature phase data signal and the second orderintermodulation estimate signal and generating a quadrature phasecross-covariance value.
 8. The system of claim 1 further comprising asecond order intermodulation estimation system receiving a receiversignal and generating a voltage envelope signal.
 9. The system of claim1 wherein the divider is a digital device.
 10. A direct conversionreceiver comprising: a second order intermodulation product correctionstage for receiving an amplified transmitted signal and a data signaland generating a correction signal from the amplified transmitted signaland the data signal; a correction system for subtracting the correctionsignal from the data signal; and wherein the second orderintermodulation product correction stage comprises an envelope detectorfor generating an envelope signal of the amplified transmitted signal.11. The system of claim 10 further comprising a signal processing stagefor extracting the data signal from the amplified transmitted signal.12. The system of claim 10 wherein the second order intermodulationproduct correction stage comprises a correlator for receiving a secondorder intermodulation estimate signal and the data signal and generatingan adaptive gain control signal.
 13. The system of claim 10 wherein thesecond order intermodulation product correction stage comprises across-covariance system for receiving the data signal and a second orderintermodulation estimate signal and generating a cross-covariance value.14. The system of claim 10 further comprising an adaptive gain unit forreceiving the second order intermodulation estimate signal and theadaptive gain control signal and generating the correction signal.
 15. Amethod for correcting a second order intermodulation product,comprising: dividing a cross-covariance signal by an auto-covariancesignal to yield an output signal; and generating a second orderintermodulation product correction factor using the output signal. 16.The method of claim 15 further comprising generating the auto-covariancesignal from a second order intermodulation product data signal.
 17. Themethod of claim 15 further comprising generating the cross-covariancesignal from a second order intermodulation product signal and a datasignal.
 18. The method of claim 15 further comprising calculating arunning average of the second order intermodulation product correctionfactor.
 19. The method of claim 15 further comprising: (a) multiplying astored second order intermodulation product correction factor by a firstweighting factor; (b) multiplying the second order intermodulationproduct correction factor by a second weighting factor; and adding theresults of steps (a) and (b) to generate an updated second orderintermodulation product correction factor.
 20. The method of claim 15further comprising storing the second order intermodulation productcorrection factor for use in calculating the second orderintermodulation product correction factor for a subsequent transmissionslot.